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precision low noise, low input bias current operational amplifiers op1177/op2177/op4177 rev. g information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. specifications subject to change without notice. no license is granted by implication or otherwise under any patent or patent rights of analog devices. trademarks and registered trademarks are the property of their respective owners. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.461.3113 ?2001C2009 analog devices, inc. all rights reserved. features low offset voltage: 60 v maximum very low offset voltage drift: 0.7 v/c maximum low input bias current: 2 na maximum low noise: 8 nv/hz typical cmrr, psrr, and a vo > 120 db minimum low supply current: 400 a per amplifier dual supply operation: 2.5 v to 15 v unity-gain stable no phase reversal inputs internally protected beyond supply voltage applications wireless base station control circuits optical network control circuits instrumentation sensors and controls thermocouples resistor thermal detectors (rtds) strain bridges shunt current measurements precision filters pin configurations ?in +in v? v+ nc nc 18 op1177 nc out nc = no connect 45 02627-001 1 2 3 4 8 7 6 5 ? in v? + in v+ out nc nc nc nc = no connect op1177 02627-002 figure 1. 8-lead msop (rm suffix) figure 2. 8-lead soic_n (r suffix) ?in a +in a v? out b +in b v+ 18 op2177 out a ?in b 45 02627-003 1 2 3 4 8 7 6 5 ?in a v? +in a out b ?in b v+ +in b out a op2177 02627-004 figure 3. 8-lead msop (rm suffix) figure 4. 8-lead soic_n (r suffix) out b 78 +in b 5 10 ?in b 69 v+ 4 11 ?in a 21 3 +in a 3 12 out a 1 14 out c +in c ?in c v? ?in d +in d out d op4177 02627-005 out a ?in a +in a v+ +in b ?in b out b ?in d +in d v? out d ?in c out c +in c 14 8 1 7 op4177 02627-006 figure 5. 14-lead soic_n (r suffix) figure 6. 14-lead tssop (ru suffix) general description the opx177 family consists of very high precision, single, dual, and quad amplifiers featuring extremely low offset voltage and drift, low input bias current, low noise, and low power consump- tion. outputs are stable with capacitive loads of over 1000 pf with no external compensation. supply current is less than 500 a per amplifier at 30 v. internal 500 series resistors protect the inputs, allowing input signal levels several volts beyond either supply without phase reversal. unlike previous high voltage amplifiers with very low offset voltages, the op1177 (single) and op2177 (dual) amplifiers are available in tiny 8-lead surface-mount msop and 8-lead narrow soic packages. the op4177 (quad) is available in tssop and 14-lead narrow soic packages. moreover, specified performance in the msop and the tssop is identical to performance in the soic package. msop and tssop are available in tape and reel only. the opx177 family offers the widest specified temperature range of any high precision amplifier in surface-mount packaging. all versions are fully specified for operation from ?40c to +125c for the most demanding operating environments. applications for these amplifiers include precision diode power measurement, voltage and current level setting, and level detection in optical and wireless transmission systems. additional applications include line-powered and portable instrumentation and controlsthermocouple, rtd, strain- bridge, and other sensor signal conditioningand precision filters.
op1177/op2177/op4177 rev. g | page 2 of 24 table of contents features .............................................................................................. 1 applications ....................................................................................... 1 pin configurations ........................................................................... 1 general description ......................................................................... 1 revision history ............................................................................... 2 specifications ..................................................................................... 3 electrical characteristics ............................................................. 4 absolute maximum ratings ............................................................ 5 thermal resistance ...................................................................... 5 esd caution .................................................................................. 5 typical performance characteristics ............................................. 6 functional description .................................................................. 14 total noise-including source resistors ................................... 14 gain linearity ............................................................................. 14 input overvoltage protection ................................................... 15 output phase reversal ............................................................... 15 settling time ............................................................................... 15 overload recovery time .......................................................... 15 thd + noise ............................................................................... 16 capacitive load drive ............................................................... 16 stray input capacitance compensation .................................. 17 reducing electromagnetic interference .................................. 17 proper board layout .................................................................. 18 difference amplifiers ................................................................ 18 a high accuracy thermocouple amplifier ........................... 19 low power linearized rtd ...................................................... 19 single operational amplifier bridge ....................................... 20 realization of active filters .......................................................... 21 band-pass krc or sallen-key filter ........................................ 21 channel separation .................................................................... 21 references on noise dynamics and flicker noise ............... 21 outline dimensions ....................................................................... 22 ordering guide .......................................................................... 24 revision history 11/09rev. f to rev. g changes to figure 64 ...................................................................... 19 changes to ordering guide .......................................................... 24 updated outline dimensions ....................................................... 22 5/09rev. e to rev. f changes to figure 64 ...................................................................... 19 changes to ordering guide .......................................................... 24 10/07rev. d to rev. e changes to general description .................................................... 1 changes to table 4 ............................................................................ 5 updated outline dimensions ....................................................... 22 7/06rev. c to rev. d changes to table 4 ............................................................................ 5 changes to figure 51 ...................................................................... 14 changes to figure 52 ...................................................................... 15 changes to figure 54 ...................................................................... 16 changes to figure 58 to figure 61 ................................................ 17 changes to figure 62 and figure 63 ............................................. 18 changes to figure 64 ...................................................................... 19 changes to figure 65 and figure 66 ............................................. 20 changes to figure 67 and figure 68............................................. 21 removed spice model section ................................................... 21 updated outline dimensions ....................................................... 22 changes to ordering guide .......................................................... 24 4/04rev. b to rev. c changes to ordering guide ............................................................. 4 changes to tpc 6 .............................................................................. 5 changes to tpc 26 ............................................................................ 7 updated outline dimensions ....................................................... 17 4/02rev. a to rev. b added op4177 ......................................................................... global edits to specifications ....................................................................... 2 edits to electrical characteristics headings .................................. 4 edits to ordering guide ................................................................... 4 11/01rev. 0 to rev. a edit to features .................................................................................. 1 edits to tpc 6 ................................................................................... 5 7/01revision 0: initial version
op1177/op2177/op4177 rev. g | page 3 of 24 specifications electrical characteristics v s = 5.0 v, v cm = 0 v, t a = 25c, unless otherwise noted. table 1. parameter symbol conditions min typ 1 max unit input characteristics offset voltage op1177 v os 15 60 v op2177/op4177 v os 15 75 v op1177/op2177 v os ?40c < t a < +125c 25 100 v op4177 v os ?40c < t a < +125c 25 120 v input bias current i b ?40c < t a < +125c ?2 +0.5 +2 na input offset current i os ?40c < t a < +125c ?1 +0.2 +1 na input voltage range ?3.5 +3.5 v common-mode rejection ratio cmrr v cm = ?3.5 v to +3.5 v 120 126 db ?40c < t a < +125c 118 125 db large signal voltage gain a vo r l = 2 k, v o = ?3.5 v to +3.5 v 1000 2000 v/mv offset voltage drift op1177/op2177 v os /t ?40c < t a < +125c 0.2 0.7 v/c op4177 v os /t ?40c < t a < +125c 0.3 0.9 v/c output characteristics output voltage high v oh i l = 1 ma, ?40c < t a < +125c +4 +4.1 v output voltage low v ol i l = 1 ma, ?40c < t a < +125c ?4.1 ?4 v output current i out v dropout < 1.2 v 10 ma power supply power supply rejection ratio op1177 psrr v s = 2.5 v to 15 v 120 130 db ?40c < t a < +125c 115 125 db op2177/op4177 psrr v s = 2.5 v to 15 v 118 121 db ?40c < t a < +125c 114 120 db supply current per amplifier i sy v o = 0 v 400 500 a ?40c < t a < +125c 500 600 a dynamic performance slew rate sr r l = 2 k 0.7 v/s gain bandwidth product gbp 1.3 mhz noise performance voltage noise e n p-p 0.1 hz to 10 hz 0.4 v p-p voltage noise density e n f = 1 khz 7.9 8.5 nv/hz current noise density i n f = 1 khz 0.2 pa/hz multiple amplifiers channel separation c s dc 0.01 v/v f = 100 khz ?120 db 1 typical values cover all parts within one standard deviation of the average value. average values given in many competitor dat a sheets as typical give unrealistically low estimates for parameters that can ha ve both positive an d negative values.
op1177/op2177/op4177 rev. g | page 4 of 24 electrical characteristics v s = 15 v, v cm = 0 v, t a = 25c, unless otherwise noted. table 2. parameter symbol conditions min typ 1 max unit input characteristics offset voltage op1177 v os 15 60 v op2177/op4177 v os 15 75 v op1177/op2177 v os ?40c < t a < +125c 25 100 v op4177 v os ?40c < t a < +125c 25 120 v input bias current i b ?40c < t a < +125c ?2 +0.5 +2 na input offset current i os ?40c < t a < +125c ?1 +0.2 +1 na input voltage range ?13.5 +13.5 v common-mode rejection ratio cmrr v cm = ?13.5 v to +13.5 v, ?40c < t a < +125c 120 125 db large signal voltage gain a vo r l = 2 k, v o = C13.5 v to +13.5 v 1000 3000 v/mv offset voltage drift op1177/op2177 v os /t ?40c < t a < +125c 0.2 0.7 v/c op4177 v os /t ?40c < t a < +125c 0.3 0.9 v/c output characteristics output voltage high v oh i l = 1 ma, ?40c < t a < +125c +14 +14.1 v output voltage low v ol i l = 1 ma, ?40c < t a < +125c ?14.1 ?14 v output current i out v dropout < 1.2 v 10 ma short-circuit current i sc 25 ma power supply power supply rejection ratio op1177 psrr v s = 2.5 v to 15 v 120 130 db ?40c < t a < +125c 115 125 db op2177/op4177 psrr v s = 2.5 v to 15 v 118 121 db ?40c < t a < +125c 114 120 db supply current per amplifier i sy v o = 0 v 400 500 a ?40c < t a < +125c 500 600 a dynamic performance slew rate sr r l = 2 k 0.7 v/s gain bandwidth product gbp 1.3 mhz noise performance voltage noise e n p-p 0.1 hz to 10 hz 0.4 v p-p voltage noise density e n f = 1 khz 7.9 8.5 nv/hz current noise density i n f = 1 khz 0.2 pa/hz multiple amplifiers channel separation c s dc 0.01 v/v f = 100 khz ?120 db 1 typical values cover all parts within one standard deviation of the average value. average values given in many competitor dat a sheets as typical give unrealistically low estimates for parameters that can ha ve both positive an d negative values.
op1177/op2177/op4177 rev. g | page 5 of 24 absolute maximum ratings table 3. parameter rating supply voltage 36 v input voltage v s? to v s+ differential input voltage supply voltage storage temperature range r, rm, and ru packages ?65c to +150c operating temperature range op1177/op2177/op4177 ?40c to +125c junction temperature range r, rm, and ru packages ?65c to +150c lead temperature, soldering (10 sec) 300c stresses above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. thermal resistance ja is specified for the worst-case conditions, that is, a device soldered in a circuit board for surface-mount packages. table 4. thermal resistance package type ja jc unit 8-lead msop (rm-8) 1 190 44 c/w 8-lead soic_n (r-8) 158 43 c/w 14-lead soic_n (r-14) 120 36 c/w 14-lead tssop (ru-14) 240 43 c/w 1 msop is available in tape and reel only. esd caution
op1177/op2177/op4177 rev. g | page 6 of 24 typical performance characteristics input offset voltage (v) number of amplifiers 45 40 35 30 25 20 15 10 5 ?30 ?20 ?10 0 10 20 30 40 0 50 ?40 v sy = 15v 02627-007 figure 7. input offset voltage distribution input offset voltage drift (v/c) number of amplifiers 80 70 60 50 40 30 20 10 0.15 0.25 0.35 0.45 0.55 0.65 0 90 0.05 v sy = 15v 02627-008 figure 8. input offset voltage drift distribution input bias current (na) number of amplifiers 120 100 80 60 40 20 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0 140 0 v sy = 15v 02627-009 figure 9. input bias current distribution load current (ma) output voltage (v) 0.01 0.1 1 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 1.8 0.001 10 source sink v sy = 15v t a = 25c 02627-010 figure 10. output voltage to supply rail vs. load current temperature (c) v sy = 15v input bias current (na) 2 1 0 ?1 ?2 0 50 100 3 ?3 ?50 150 0 2627-011 figure 11. input bias current vs. temperature frequency (hz) phase shift (degrees) open-loop gain (db) 1m 50 40 30 20 10 0 ?10 60 ?20 100k 10m 225 180 135 90 45 0 ?45 270 ?90 gain phase v sy = 15v c l = 0 r l = 02627-012 figure 12. open-loop gain and phase shift vs. frequency
op1177/op2177/op4177 rev. g | page 7 of 24 frequency (hz) closed-loop gain (db) 10k 100k 1m 10m 100 80 60 40 20 0 ?20 ?40 ?60 120 ?80 1k 100m v sy = 15v v in = 4mv p-p c l = 0 r l = a v = 100 a v = 1 a v = 10 02627-013 figure 13. closed-loop gain vs. frequency frequency (hz) output impedance ( ? ) 1k 10k 100k 1m 450 400 350 300 250 200 150 100 50 100 500 0 v sy = 15v v in = 50mv p-p a v = 10 a v = 100 a v = 1 0 2627-014 figure 14. output im pedance vs. frequency time (100s/div) voltage (1v/div) gnd v sy = 15v c l = 300pf r l = 2k ? v in = 4v a v = 1 02627-015 figure 15. large signal transient response time (100s/div) voltage (100mv/div) gnd v sy = 15v c l = 1,000pf r l = 2k ? v in = 100mv a v = 1 02627-016 figure 16. small signal transient response capacitance (pf) small signal overshoot (%) 10 100 1k 1 10k 45 40 35 30 25 20 15 10 5 50 0 +os ?os v sy = 15v r l = 2k ? v in = 100mv p-p 02627-017 figure 17. small signal overshoot vs. load capacitance time (10s/div) +200m v 0v ?15v 0v v sy = 15v r l = 10k ? a v = ?100 v in = 200mv input output 0 2627-018 figure 18. positive overvoltage recovery
op1177/op2177/op4177 rev. g | page 8 of 24 time (4s/div) 0v ? 200m v 0v 15v v sy = 15v r l = 10k ? a v = ?100 v in = 200mv input output 0 2627-019 figure 19. negative overvoltage recovery frequency (hz) cmrr (db) 100 1k 10k 100k 1m 120 100 80 60 40 20 140 0 10 10m v sy = 15v 0 2627-020 figure 20. cmrr vs. frequency frequency (hz) psrr (db) 100 1k 10k 100k 1m 120 100 80 60 40 20 140 0 10 10m v sy = 15v +psrr ?psrr 02627-021 figure 21. psrr vs. frequency v noise (0.2v/div) time (1s/div) v sy = 15v 0 2627-022 figure 22. 0.1 hz to 10 hz input voltage noise frequency (hz) 16 14 12 10 8 6 4 18 2 50 100 150 200 0 250 v sy = 15v voltage noise density (nv/ hz) 02627-023 figure 23. voltage noise density vs. frequency short-circuit current (ma) +i sc ?i sc temperature (c) 30 25 20 15 10 5 0 50 100 35 0 ?50 150 v sy = 15v 0 2627-024 figure 24. short-circuit current vs. temperature
op1177/op2177/op4177 rev. g | page 9 of 24 output voltage swing (v) 14.40 14.00 14.30 14.05 14.25 14.20 14.15 14.10 14.35 +v oh ?v ol temperature (c) 0 50 100 ?50 150 v sy = 15v 02627-025 figure 25. output voltag e swing vs. temperature time from power supply turn-on (sec) offset voltage (v) 0.4 0.3 0.2 0.1 0 ?0.1 ?0.2 ?0.3 0.5 ?0.5 ?0.4 20 40 60 80 100 120 0 140 v sy = 15v 02627-026 figure 26. warm-up drift input offset voltage (v) 0 12 8 4 14 18 2 6 10 16 temperature (c) v sy = 15v 0 50 100 ?50 150 02627-027 figure 27. input offset voltage vs. temperature cmrr (db) 123 127 125 128 129 124 126 130 131 132 133 temperature (c) v sy = 15v 0 50 100 ?50 150 0 2627-028 figure 28. cmrr vs. temperature psrr (db) 123 127 125 128 129 124 126 130 131 132 133 temperature (c) v sy = 15v 0 50 100 ?50 150 02627-029 figure 29. psrr vs. temperature input offset voltage (v) number of amplifiers 50 15 0 45 20 10 5 30 25 40 35 v sy = 5v ?40 ?30 ?20 ?10 0 10 20 30 40 0 2627-030 figure 30. input offset voltage distribution
op1177/op2177/op4177 rev. g | page 10 of 24 output voltage (v) load current (ma) 1.4 0.8 0 0.4 0.2 0.6 1.0 1.2 0.01 0.1 1 v sy = 5v t a = 25c sink source 0.001 10 02627-031 figure 31. output voltage to supply rail vs. load current 0 2627-032 frequency (hz) phase shift (degrees) open-loop gain (db) 1m 50 40 30 20 10 0 ?10 60 ?20 100k 10m 225 180 135 90 45 0 ?45 270 ?90 gain phase v sy = 5v c l = 0 r l = figure 32. open-loop gain and phase shift vs. frequency frequency (hz) closed-loop gain (db) 10k 100k 1m 10m 100 80 60 40 20 0 ?20 ?40 ?60 120 ?80 1k 100m v sy = 5v v in = 4mv p-p c l = 0 r l = a v = 100 a v = 1 a v = 10 0 2627-033 figure 33. closed-loop gain vs. frequency frequency (hz) output impedance ( ? ) 1k 10k 100k 100 1m 450 400 350 300 250 200 150 100 50 500 0 v sy = 5v v in = 50mv p-p a v = 100 a v = 1 a v = 10 02627-034 figure 34. output im pedance vs. frequency time (100s/div) voltage (1v/div) gnd v sy = 5v c l = 300pf r l = 2k ? v in = 1v a v = 1 02627-035 figure 35. large signal transient response time (10s/div) voltage (50mv/div) gnd v sy = 5v c l = 1,000pf r l = 2k ? v in = 100mv a v = 1 02627-036 figure 36. small signal transient response
op1177/op2177/op4177 rev. g | page 11 of 24 capacitance (pf) small signal overshoot (%) 10 100 1k 1 10k 45 40 35 30 25 20 15 10 5 50 0 +os ?os v sy = 5v r l = 2k ? v in = 100mv 02627-037 figure 37. small signal overshoot vs. load capacitance time (4s/div) +200mv 0v ?15v 0v v sy = 5v r l = 10k ? a v = ?100 v in = 200mv input output 02627-038 figure 38. positive overvoltage recovery time (4s/div) 0v ? 200m v 0v 5v v sy = 5v r l = 10k ? a v = ?100 v in = 200mv input output 02627-039 figure 39. negative overvoltage recovery time (200s/div) voltage (2v/div) gnd v s = 5v a v = 1 r l = 10k ? input output 02627-040 figure 40. no phase reversal frequency (hz) cmrr (db) 100 1k 10k 100k 1m 120 100 80 60 40 20 140 0 10 10m v sy = 5v 02627-041 figure 41. cmrr vs. frequency frequency (hz) psrr (db) 100 1k 10k 100k 1m 160 120 80 40 200 0 10 10m v sy = 5v 140 100 60 20 180 +psrr ?psrr 02627-042 figure 42. psrr vs. frequency
op1177/op2177/op4177 rev. g | page 12 of 24 v noise (0.2v/div) time (1s/div) v sy = 5v 02627-043 figure 43. 0.1 hz to 10 hz input voltage noise frequency (hz) 16 14 12 10 8 6 4 18 2 50 100 150 200 02 5 0 v sy = 5v voltage noise density (nv/ hz) 02627-044 figure 44. voltage noise density vs. frequency short-circuit current (ma) +i sc ?i sc temperature (c) 30 25 20 15 10 5 0 50 100 35 0 ?50 150 v sy = 5v 02627-045 figure 45. short-circuit current vs. temperature output voltage swing (v) 4.40 4.00 4.30 4.05 4.25 4.20 4.15 4.10 4.35 +v oh ?v ol temperature (c) 0 50 100 ?50 150 v sy = 5v 02627-046 figure 46. output voltag e swing vs. temperature input offset voltage (v) 0 10 5 20 25 15 temperature (c) v sy = 5v 0 50 100 ?50 150 02627-047 figure 47. input offset voltage vs. temperature supply current (a) 0 300 200 500 600 400 temperature (c) 0 50 100 ?50 150 100 v sy = 5v v sy = 15v 02627-048 figure 48. supply current vs. temperature
op1177/op2177/op4177 rev. g | page 13 of 24 supply current (a) 0 300 200 100 350 450 50 150 250 400 supply voltage (v) 51015 02 0 25 30 35 t a = 25c 0 2627-049 figure 49. supply current vs. supply voltage frequency (hz) channel separation (db) 100 1k 10k 100k ?20 ?40 ?60 ?80 ?100 ?120 ?140 0 ?160 10 1m 02627-050 figure 50. channel separation vs. frequency
op1177/op2177/op4177 rev. g | page 14 of 24 functional description the opx177 series is the fourth generation of analog devices, inc., industry-standard op07 amplifier family. opx177 is a high precision, low noise operational amplifier with a combination of extremely low offset voltage and very low input bias currents. unlike jfet amplifiers, the low bias and offset currents are relatively insensitive to ambient temperatures, even up to 125c. analog devices proprietary process technology and linear design expertise has produced a high voltage amplifier with superior performance to the op07 , op77 , and op177 in a tiny msop 8-lead package. despite its small size, the opx177 offers numerous improvements, including low wideband noise, very wide input and output voltage range, lower input bias current, and complete freedom from phase inversion. opx177 has a specified operating temperature range as wide as any similar device in a plastic surface-mount package. this is increasingly important as pcb and overall system sizes continue to shrink, causing internal system temperatures to rise. power consumption is reduced by a factor of four from the op177, and bandwidth and slew rate increase by a factor of two. the low power dissipation and very stable performance vs. temperature also act to reduce warmup drift errors to insignificant levels. open-loop gain linearity under heavy loads is superior to compet- itive parts, such as the opa277, improving dc accuracy and reducing distortion in circuits with high closed-loop gains. inputs are internally protected from overvoltage conditions referenced to either supply rail. like any high performance amplifier, maximum performance is achieved by following appropriate circuit and pcb guidelines. the following sections provide practical advice on getting the most out of the opx177 under a variety of application conditions. total noise-including source resistors the low input current noise and input bias current of the opx177 make it useful for circuits with substantial input source resistance. input offset voltage increases by less than 1 v maximum per 500 of source resistance. the total noise density of the opx177 is () s s nn totaln ktrriee 4 2 2 , ++= where: e n is the input voltage noise density. i n is the input current noise density. r s is the source resistance at the noninverting terminal. k is boltzmanns constant (1.38 10 ?23 j/k). t is the ambient temperature in kelvin ( t = 273 + temperature in degrees celsius). for r s < 3.9 k, e n dominates and e n,total e n for 3.9 k < r s < 412 k, voltage noise of the amplifier, the current noise of the amplifier translated through the source resistor, and the thermal noise from the source resistor all contribute to the total noise. for r s > 412 k, the current noise dominates and e n,total i n r s the total equivalent rms noise over a specific bandwidth is expressed as ( ) bw ee totaln n , = where bw is the bandwidth in hertz. the preceding analysis is valid for frequencies larger than 50 hz. when considering lower frequencies, flicker noise (also known as 1/f noise) must be taken into account. for a reference on noise calculations, refer to the band-pass krc or sallen-key filter section. gain linearity gain linearity reduces errors in closed-loop configurations. the straighter the gain curve, the lower the maximum error over the input signal range. this is especially true for circuits with high closed-loop gains. the op1177 has excellent gain line arity even with heavy loads, as shown in figure 51 . compare its performance to the opa277, shown in figure 52 . both devices are measured under identical conditions, with r l = 2 k. the op2177 (dual) has virtually no distortion at lower voltages. compared to the opa277 at several supply voltages and various loads, op1177 performance far exceeds that of its counterpart. (5v/div) op1177 (10v/div) v s = 15v r l = 2k 02627-051 figure 51. gain linearity
op1177/op2177/op4177 rev. g | page 15 of 24 (5v/div) opa277 v sy = 15v r l = 2k ? (10v/div) 0 2627-052 figure 52. gain linearity input overvoltage protection when input voltages exceed the positive or negative supply voltage, most amplifiers require external resistors to protect them from damage. the opx177 has internal protective circuitry that allows voltages as high as 2.5 v beyond the supplies to be applied at the input of either terminal without any harmful effects. use an additional resistor in series with the inputs if the voltage exceeds the supplies by more than 2.5 v. the value of the resistor can be determined from the formula ma5 500 d : s s in r vv with the opx177 low input offset current of <1 na maximum, placing a 5 k resistor in series with both inputs adds less than 5 v to input offset voltage and has a negligible impact on the overall noise performance of the circuit. 5 k protects the inputs to more than 27 v beyond either supply. refer to the thd + noise section for additional information on noise vs. source resistance. output phase reversal phase reversal is defined as a change of polarity in the amplifier transfer function. many operational amplifiers exhibit phase reversal when the voltage applied to the input is greater than the maximum common-mode voltage. in some instances, this can cause permanent damage to the amplifier. in feedback loops, it can result in system lockups or equipment damage. the opx177 is immune to phase reversal problems even at input voltages beyond the supplies. v sy = 10v a v = 1 time (400s/div) v in v out voltage (5v/div) 02627-053 figure 53. no phase reversal settling time settling time is defined as the time it takes an amplifier output to reach and remain within a percentage of its final value after application of an input pulse. it is especially important in measure- ment and control circuits in which amplifiers buffer adc inputs or dac outputs. to minimize settling time in amplifier circuits, use proper bypassing of power supplies and an appropriate choice of circuit components. resistors should be metal film types, because they have less stray capacitance and inductance than their wire-wound counterparts. capacitors should be polystyrene or polycarbonate types to minimize dielectric absorption. the leads from the power supply should be kept as short as possible to minimize capacitance and inductance. the opx177 has a settling time of about 45 s to 0.01% (1 mv) with a 10 v step applied to the input in a noninverting unity gain. overload recovery time overload recovery is defined as the time it takes the output voltage of an amplifier to recover from a saturated condition to its linear response region. a common example is one in which the output voltage demanded by the transfer function of the circuit lies beyond the maximum output voltage capability of the amplifier. a 10 v input applied to an amplifier in a closed- loop gain of 2 demands an output voltage of 20 v. this is beyond the output voltage range of the opx177 when operating at 15 v supplies and forces the output into saturation. recovery time is important in many applications, particularly where the operational amplifier must amplify small signals in the presence of large transient voltages.
op1177/op2177/op4177 rev. g | page 16 of 24 op1177 6 7 2 3 4 v+ v? r2 100k ? v out 10k ? r1 1k? + ? 2 00m v 02627-054 figure 54. test circuit for overload recovery time figure 18 shows the positive overload recovery time of the op1177. the output recovers in less than 4 s after being overdriven by more than 100%. the negative overload recovery of the op1177 is 1.4 s, as seen in figure 19 . thd + noise the opx177 has very low total harmonic distortion. this indicates excellent gain linearity and makes the opx177 a great choice for high closed-loop gain precision circuits. figure 55 shows that the opx177 has approximately 0.00025% distortion in unity gain, the worst-case configuration for distortion. frequency (hz) thd + n (%) 100 1k 0.001 0.01 20 6k 0.0001 0.1 02627-055 v sy = 15v r l = 10k ? bw = 22khz figure 55. thd + n vs. frequency capacitive load drive opx177 is inherently stable at all gains and capable of driving large capacitive loads without oscillation. with no external compensation, the opx177 safely drives capacitive loads up to 1000 pf in any configuration. as with virtually any amplifier, driving larger capacitive loads in unity gain requires additional circuitry to assure stability. in this case, a snubber network is used to prevent oscillation and reduce the amount of overshoot. a significant advantage of this method is that it does not reduce the output swing because the resistor r s is not inside the feedback loop. figure 56 is a scope shot of the output of the opx177 in response to a 400 mv pulse. the load capacitance is 2 nf. the circuit is configured in positive unity gain, the worst-case condition for stability. as shown in figure 58 , placing an r-c network parallel to the load capacitance (c l ) allows the amplifier to drive higher values of c l without causing oscillation or excessive overshoot. there is no ringing, and overshoot is reduced from 27% to 5% using the snubber network. optimum values for r s and c s are tabulated in table 5 for several capacitive loads, up to 200 nf. values for other capacitive loads can be determined experimentally. table 5. optimum values for capacitive loads c l r s c s 10 nf 20 0.33 f 50 nf 30 6.8 nf 200 nf 200 0.47 f 0 gnd voltage (200mv/div) time (10s/div) v sy = 5v r l = 10k ? c l = 2nf 0 2627-056 figure 56. capacitive load drive without snubber gnd vol t age (200mv/div) time (10s/div) v sy = 5v r l = 10k ? r s = 200 ? c l = 2nf c s = 0.47f 0 2627-057 figure 57. capacitive load drive with snubber
op1177/op2177/op4177 rev. g | page 17 of 24 op1177 6 7 2 3 4 v+ v? v out r s + ? 400mv c s c l 0 2627-058 figure 58. snubber network configuration caution: the snubber technique cannot recover the loss of bandwidth induced by large capacitive loads. stray input capacitance compensation the effective input capacitance in an operational amplifier circuit (c t ) consists of three components. these are the internal differential capacitance between the input terminals, the internal common-mode capacitance of each input to ground, and the external capacitance including parasitic capacitance. in the circuit in figure 59 , the closed-loop gain increases as the signal frequency increases. the transfer function of the circuit is () r1sc r1 r2 t ++ 1 1 indicating a zero at () t t cr2r1 r2r1c r1r2 s / 2 1 = + = depending on the value of r1 and r2, the cutoff frequency of the closed-loop gain can be well below the crossover frequency. in this case, the phase margin ( m ) can be severely degraded, resulting in excessive ringing or even oscillation. a simple way to overcome this problem is to insert a capacitor in the feedback path, as shown in figure 60 . the resulting pole can be positioned to adjust the phase margin. setting c f = (r1/r2) c t achieves a phase margin of 90. r2 r1 v1 + ? op1177 2 3 v out c t 02627-059 6 7 4 v+ v? figure 59. stray input capacitance r2 r1 v1 + ? op1177 2 3 v out c t c f 02627-060 6 7 4 v+ v? figure 60. compensation using feedback capacitor reducing electromagnetic interference a number of methods can be utilized to reduce the effects of emi on amplifier circuits. in one method, stray signals on either input are coupled to the opposite input of the amplifier. the result is that the signal is rejected according to the cmrr of the amplifier. this is usually achieved by inserting a capacitor between the inputs of the amplifier, as shown in figure 61 . however, this method can also cause instability, depending on the value of capacitance. r2 r1 v1 + ? op1177 2 3 v out c 02627-061 6 7 4 v+ v? figure 61. emi reduction placing a resistor in series with the capacitor (see figure 62 ) increases the dc loop gain and reduces the output error. positioning the breakpoint (introduced by r-c) below the secondary pole of the operational amplifier improves the phase margin and, therefore, stability. r can be chosen independently of c for a specific phase margin according to the formula () ? ? ? ? ? ? +?= r1 r2 jfa r2 r 2 1 where: a is the open-loop gain of the amplifier. f 2 is the frequency at which the phase of a = m ? 180. op1177 2 3 r c r1 r2 v out v1 + ? 02627-062 6 7 4 v+ v? figure 62. compensation using input r-c network
op1177/op2177/op4177 rev. g | page 18 of 24 proper board layout the opx177 is a high precision device. to ensure optimum performance at the pcb level, care must be taken in the design of the board layout. to avoid leakage currents, the surface of the board should be kept clean and free of moisture. coating the surface creates a barrier to moisture accumulation and helps reduce parasitic resistance on the board. keeping supply traces short and properly bypassing the power supplies minimizes power supply disturbances due to output current variation, such as when driving an ac signal into a heavy load. bypass capacitors should be connected as closely as possible to the device supply pins. stray capacitances are a concern at the outputs and the inputs of the amplifier. it is recommended that signal traces be kept at least 5 mm from supply lines to minimize coupling. a variation in temperature across the pcb can cause a mismatch in the seebeck voltages at solder joints and other points where dissi- milar metals are in contact, resulting in thermal voltage errors. to minimize these thermocouple effects, orient resistors so heat sources warm both ends equally. input signal paths should contain matching numbers and types of components, where possible to match the number and type of thermocouple junctions. for example, dummy components such as zero value resistors can be used to match real resistors in the opposite input path. matching components should be located in close proximity and should be oriented in the same manner. ensure leads are of equal length so that thermal conduction is in equilibrium. keep heat sources on the pcb as far away from amplifier input circuitry as is practical. the use of a ground plane is highly recommended. a ground plane reduces emi noise and also helps to maintain a constant temperature across the circuit board. difference amplifiers difference amplifiers are used in high accuracy circuits to improve the common-mode rejection ratio (cmrr). r1 v 1 v 2 r3 = r1 r4 = r1 op1177 2 3 = r4 r3 r2 r1 r2 100k? v out 02627-063 6 7 4 v+ v? figure 63. difference amplifier in the single instrumentation amplifier (see figure 63 ), where r1 r2 r3 r4 12 o vv r1 r2 v a mismatch between the ratio r2/r1 and r4/r3 causes the common-mode rejection ratio to be reduced. to better understand this effect, consider that, by definition, cm dm a a cmrr where adm is the differential gain and acm is the common- mode gain. cm o cm diff o dm v v a v v a and 21 cm 21 diff vvvvvv 2 1 and for this circuit to act as a difference amplifier, its output must be proportional to the differential input signal. from figure 63 , 2 1 o v r4 r3 r1 r2 v r1 r2 v 1 1 arranging terms and combining the previous equations yields r2r3 r4r1 r4r2 r3r2 r4r1 cmrr 22 2 (1) the sensitivity of cmrr with respect to the r1 is obtained by taking the derivative of cmrr, in equation 1, with respect to r1. g g g g r2r3 r1r4 r2r3 r2r4 r2r3 r1r4 r1r4 r1r1 cmrr 22 2 22 r1r4 r2r3 r1 cmrr 2 2 1 g g assuming that r1 r2 r3 r4 r and r (1 ? ) < r1 , r2 , r3 , r4 < r (1 + ) the worst-case cmrr error arises when r1 = r4 = r (1 + ) and r2 = r3 = r (1 ? )
op1177/op2177/op4177 rev. g | page 19 of 24 plugging these values into equation 1 yields ? 2 1 min cmrr where is the tolerance of the resistors. lower tolerance value resistors result in higher common-mode rejection (up to the cmrr of the operational amplifier). using 5% tolerance resistors, the highest cmrr that can be guaranteed is 20 db. alternatively, using 0.1% tolerance resistors results in a common-mode rejection ratio of at least 54 db (assuming that the operational amplifier cmrr 54 db). with the cmrr of opx177 at 120 db minimum, the resistor match is the limiting factor in most circuits. a trimming resistor can be used to further improve resistor matching and cmrr of the difference amplifier circuit. a high accuracy thermocouple amplifier a thermocouple consists of two dissimilar metal wires placed in contact. the dissimilar metals produce a voltage v tc = (t j ? t r ) where: t j is the temperature at the measurement of the hot junction. t r is the temperature at the cold junction. is the seebeck coefficient specific to the dissimilar metals used in the thermocouple. v tc is the thermocouple voltage and becomes larger with increasing temperature. maximum measurement accuracy requires cold junction compen- sation of the thermocouple. to perform the cold junction compen- sation, apply a copper wire short across the terminating junctions (inside the isothermal block) simulating a 0c point. adjust the output voltage to zero using the r5 trimming resistor, and remove the copper wire. the opx177 is an ideal amplifier for thermocouple circuits because it has a very low offset voltage, excellent psrr and cmrr, and low noise at low frequencies. it can be used to create a thermocouple circuit with great linearity. resistor r1, resistor r2, and diode d1, shown in figure 64 , are mounted in an isothermal block. v+ 7 4 cu cu tr tr d1 d1 adr293 v cc c1 2.2f r3 47k ? 10f r2 4.02k ? r8 1k ? r7 80.6k ? r6 50 ? r9 200k ? 0.1f 10f 0.1f 10f v? 10f r4 50 ? r5 100 ? r1 50 ? isothermal block v tc t j (?) (+) 6 2 3 op1177 v out 02627-064 figure 64. type k thermocouple amplifier circuit low power linearized rtd a common application for a single element varying bridge is an rtd thermometer amplifier, as shown in figure 65 . the excita- tion is delivered to the bridge by a 2.5 v reference applied at the top of the bridge. rtds may have thermal resistance as high as 0.5c to 0.8c per mw. to minimize errors due to resistor drift, the current through each leg of the bridge must be kept low. in this circuit, the amplifier supply current flows through the bridge. however, at the opx177 maximum supply current of 600 a, the rtd dissipates less than 0.1 mw of power, even at the highest resis- tance. errors due to power dissipation in the bridge are kept under 0.1c. calibration of the bridge is made at the minimum value of temperature to be measured by adjusting r p until the output is zero. to calibrate the output span, set the full-scale and linearity potentiometers to midpoint and apply a 500c temperature to the sensor or substitute the equivalent 500c rtd resistance. adjust the full-scale potentiometer for a 5 v output. finally, apply 250c or the equivalent rtd resistance and adjust the linearity potentiometer for 2.5 v output. the circuit achieves better than 0.5c accuracy after adjustment.
op1177/op2177/op4177 rev. g | page 20 of 24 0 2627-065 200 ? 500 ? 4.37k ? 100? 100? 20? 4.12k ? 4.12k ? 5k? 49.9k ? adr421 +15 v 0.1 f v+ 100 ? rtd 1/2 op2177 7 6 5 1/2 op2177 1 8 2 3 4 v? v ou t v out where = r / r is the fractional deviation of the rtd resistance with respect to the bridge resistance due to the change in temper- ature at the rtd. for << 1, the preceding expression becomes g g ref ref o v r2 r1 r2 r1 r r2 r2 r1 r r1 v r r2 v 1 1 with v ref constant, the output voltage is linearly proportional to with a gain factor of r2 r1 r2 r1 r r2 v ref 1 02627-066 r r r r(1+ ) adr421 15v 0.1 f op1177 6 7 4 2 3 v+ v? r f r f v out figure 65. low power linearized rtd circuit single operational amplifier bridge the low input offset voltage drift of the op1177 makes it very effective for bridge amplifier circuits used in rtd signal condi- tioning. it is often more economical to use a single bridge operational amplifier as opposed to an instrumentation amplifier. in the circuit shown in figure 66 , the output voltage at the operational amplifier is figure 66. single bridge amplifier g g 1 1 r2 r1 r r1 v r r2 v ref o
op1177/op2177/op4177 rev. g | page 21 of 24 realization of active filters band-pass krc or sallen-key filter the low offset voltage and the high cmrr of the opx177 make it an excellent choice for precision filters, such as the band-pass krc filter shown in figure 67 . this filter type offers the capability to tune the gain and the cutoff frequency independently. because the common-mode voltage into the amplifier varies with the input signal in the krc filter circuit, a high cmrr is required to minimize distortion. also, the low offset voltage of the opx177 allows a wider dynamic range when the circuit gain is chosen to be high. the circuit of figure 67 consists of two stages. the first stage is a simple high-pass filter where the corner frequency (f c ) is c1c2r1r2 2 1 (2) and r2 r1 kq = (3) where k is the dc gain. choosing equal capacitor values minimizes the sensitivity and simplifies equation 2 to r1r2c 2 1 the value of q determines the peaking of the gain vs. frequency (ringing in transient response). commonly chosen values for q are generally near unity. setting 2 1 q = yields minimum gain peaking and minimum ringing. determine values for r1 and r2 by using equation 3. for , 2 1 q = r1/r2 = 2 in the circuit example. select r1 = 5 k and r2 = 10 k for simplicity. the second stage is a low-pass filter where the corner frequency can be determined in a similar fashion. for r3 = r4 = r c4 c3 q c4 c3 r f c 2 1 and 2 1 = = channel separation multiple amplifiers on a single die are often required to reject any signals originating from the inputs or outputs of adjacent channels. op2177 input and bias circuitry is designed to prevent feedthrough of signals from one amplifier channel to the other. as a result, the op2177 has an impressive channel separation of greater than ?120 db for frequencies up to 100 khz and greater than ?115 db for signals up to 1 mhz. 02627-067 c3 680pf 1/2 op2177 7 8 6 5 4 v+ v 1/2 op2177 1 2 3 r2 10k v1 + r1 20k c2 10nf c1 10nf r3 33k r4 33k c4 330pf v out figure 67. two-stage, band-pass krc filter 1 2 3 1/2 op2177 100 10k 1/2 op2177 7 8 6 5 4 v+ v + 0 2627-068 v1 50mv figure 68. channel separation test circuit references on noise dynamics and flicker noise s. franco, design with operational amplifiers and analog integrated circuits . mcgraw-hill, 1998. analog devices, inc., the best of analog dialogue , 1967 to 1991 . analog devices, inc., 1991.
op1177/op2177/op4177 rev. g | page 22 of 24 outline dimensions controlling dimensions are in millimeters; inch dimensions (in parentheses) are rounded-off millimeter equivalents for reference only and are not appropriate for use in design. compliant to jedec standards ms-012-a a 012407-a 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) 0.50 (0.0196) 0.25 (0.0099) 45 8 0 1.75 (0.0688) 1.35 (0.0532) seating plane 0.25 (0.0098) 0.10 (0.0040) 4 1 85 5.00 (0.1968) 4.80 (0.1890) 4.00 (0.1574) 3.80 (0.1497) 1.27 (0.0500) bsc 6.20 (0.2441) 5.80 (0.2284) 0.51 (0.0201) 0.31 (0.0122) coplanarity 0.10 figure 69. 8-lead standard small outline package [soic_n] narrow body (r-8) dimensions shown in millimeters and (inches) controlling dimensions are in millimeters; inch dimensions (in parentheses) are rounded-off millimeter equivalents for reference only and are not appropriate for use in design. compliant to jedec standards ms-012-ab 060606-a 14 8 7 1 6.20 (0.2441) 5.80 (0.2283) 4.00 (0.1575) 3.80 (0.1496) 8.75 (0.3445) 8.55 (0.3366) 1.27 (0.0500) bsc seating plane 0.25 (0.0098) 0.10 (0.0039) 0.51 (0.0201) 0.31 (0.0122) 1.75 (0.0689) 1.35 (0.0531) 0.50 (0.0197) 0.25 (0.0098) 1.27 (0.0500) 0.40 (0.0157) 0.25 (0.0098) 0.17 (0.0067) coplanarity 0.10 8 0 45 figure 70. 14-lead standard small outline package [soic_n] narrow body (r-14) dimensions shown in millimeters and (inches)
op1177/op2177/op4177 rev. g | page 23 of 24 compliant to jedec standards mo-187-aa 100709-b 6 0 0.80 0.55 0.40 4 8 1 5 0.65 bsc 0.40 0.25 1.10 max 3.20 3.00 2.80 coplanarity 0.10 0.23 0.09 3.20 3.00 2.80 5.15 4.90 4.65 pin 1 identifier 15 max 0.95 0.85 0.75 0.15 0.05 figure 71. 8-lead mini small outline package [msop] (rm-8) dimensions shown in millimeters compliant to jedec standards mo-153-ab-1 061908-a 8 0 4.50 4.40 4.30 14 8 7 1 6.40 bsc pin 1 5.10 5.00 4.90 0.65 bsc 0.15 0.05 0.30 0.19 1.20 max 1.05 1.00 0.80 0.20 0.09 0.75 0.60 0.45 coplanarity 0.10 seating plane figure 72. 14-lead thin shrink small outline package [tssop] (ru-14) dimensions shown in millimeters
op1177/op2177/op4177 rev. g | page 24 of 24 ordering guide model temperature range package de scription package option branding op1177ar ?40c to +125c 8-lead soic_n r-8 op1177arz 1 ?40c to +125c 8-lead soic_n r-8 op1177arz-reel 1 ?40c to +125c 8-lead soic_n r-8 op1177arz-reel7 1 ?40c to +125c 8-lead soic_n r-8 op1177arm-reel ?40c to +125c 8-lead msop rm-8 aza op1177armz 1 ?40c to +125c 8-lead msop rm-8 aza# op1177armz-reel 1 ?40c to +125c 8-lead msop rm-8 aza# op1177armz-r7 1 ?40c to +125c 8-lead msop rm-8 aza# op2177ar ?40c to +125c 8-lead soic_n r-8 op2177ar-reel ?40c to +125c 8-lead soic_n r-8 op2177ar-reel7 ?40c to +125c 8-lead soic_n r-8 op2177arz 1 ?40c to +125c 8-lead soic_n r-8 op2177arz-reel 1 ?40c to +125c 8-lead soic_n r-8 op2177arz-reel7 1 ?40c to +125c 8-lead soic_n r-8 op2177arm-reel ?40c to +125c 8-lead msop rm-8 b2a op2177armz 1 ?40c to +125c 8-lead msop rm-8 b2a# op2177armz-reel 1 ?40c to +125c 8-lead msop rm-8 b2a# op2177armz-r7 1 ?40c to +125c 8-lead msop rm-8 b2a# op4177ar ?40c to +125c 14-lead soic_n r-14 op4177ar-reel ?40c to +125c 14-lead soic_n r-14 op4177ar-reel7 ?40c to +125c 14-lead soic_n r-14 op4177arz 1 ?40c to +125c 14-lead soic_n r-14 op4177arz-reel 1 ?40c to +125c 14-lead soic_n r-14 op4177arz-reel7 1 ?40c to +125c 14-lead soic_n r-14 op4177aru ?40c to +125c 14-lead tssop ru-14 op4177aru-reel ?40c to +125c 14-lead tssop ru-14 op4177aruz 1 ?40c to +125c 14-lead tssop ru-14 op4177aruz-reel 1 ?40c to +125c 14-lead tssop ru-14 1 z = rohs compliant part; # denotes pb-free product may be top or bottom marked. ?2001C2009 analog devices, inc. all rights reserved. trademarks and registered trademarks are the prop erty of their respective owners. d02627-0-11/09(g)

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